This invention relates to an electrical switching power supply, and in particular to a power supply, such as for use in connection with electrostatic headphones, capable of delivering high voltages without great loss of power in the switching operation.
Certain applications, such as electrostatic headphones, require a high DC polarization bias and large voltage swings from excitation amplifiers, on the order of .+-.600 volts for example. Particularly where it is desired to use batteries as the original power source, it is known to use a switching power supply to provide these voltages. Such switching power supplies are generally available in two types, those that employ pulse width modulation (PWM) and resonant types.
PWM power supplies regulate their outputs by fixing the control frequency and varying the on/off time ratio of their power switches, whether junction transistor or FET, in which case the power switch must turn off at high current or voltage conditions, dissipating a significant amount of power during the switching transition. Where the power is limited, being obtained from batteries, such power dissipation raises the overall power consumption beyond an acceptable level. The large ratio of output to input voltage further aggravates the problems of a PWM switching power supply. Capacitance on the load side of the power transformer would be reflected to the primary by the square of the turns ratio. The total capacitance, then, including that of the transformer and rectifier, would impose a highly reactive load on the power switch. The higher currents associated with such a load tend to further increase switching losses.
Most resonant mode switching power supplies achieve regulation by varying the switching frequency and fixing either the on time or the off time of the switch. FIG. 1 shows a conventional parallel resonant mode switching circuit 2. In this type of circuit, again stray and parasitic capacitances of the load side are reflected into the primary as the square of the turns ratio. As there shown, L1 is the equivalent primary inductance of the power converter transformer and capacitor C1, shown in phantom, represents the reflected load capacitance together with all other parasitic capacitance. Capacitor C2 is a bypass capacitor. Capacitance C1 and inductor L1 together form the resonant tank circuit 4. The power switch S1 directs power into the tank at zero voltage, or zero current in a series resonant tank, of the sinusoidal tank waveform. At these points, no power would be dissipated through the switch, and thus the switching losses of a resonant mode power supply would be much less than those of a PWM power supply. Unlike PWM power supplies, resonant mode power supplies oscillate at a preferred resonant frequency. At resonance, the tank is seen as a resistive load. By minimizing nonproductive reactive currents, switching losses due to these currents can be minimized.
The problem with existing resonant mode IC regulators has been that such components are generally optimized for high power applications which draw power from a conventional AC power line, meaning that the energy supply is inexhaustible, practically speaking, and energy dissipation is generally reduced merely to reduce heat production. In some cases these resonant mode IC regulators have internal circuitry that requires operating voltages greatly exceeding conventional battery voltage levels. In addition, while the power consumption of the internal circuitry in its quiescent or latent state is inconsequential when dealing with power converters connected to the AC line, it becomes very significant in a low power application such as a battery-driven headphone.
Another compromise made by designers of resonant mode power supplies in the past was that, while either the "on" time or the "off" time was fixed, the other period was permitted to vary. Which period is fixed is determined by whether the circuit is a parallel or series resonant tank arrangement. As shown in FIGS. 2A and 2B, though, regardless of which period is fixed, such an arrangement can result in poor synchronization of the switching occurrence with the zero signal occurrence. That is, if the "on" time is too short, as illustrated in FIG. 2A, power switching takes place too early. Similarly, if the "on" time is too long, as illustrated in FIG. 2B, a negative voltage tail occurs. These variations can be caused by variations in input voltage, load and other conditions. In any event, switch current and voltage are both non-zero at the moment of switching.
In the case of the voltage tail shown in FIG. 2B, most practical switching devices cannot accommodate such an occurrence. This leads designers to modify the circuit as shown in FIG. 3 to include clamp means, such as diode D1, to prevent reverse voltage or current in the circuit, resulting in the waveform shown in FIG. 2C. Although energy that is not transferred from the tank into the load is returned to the input power source through this diode D1, energy is still wasted in pushing this return current through the forward voltage drop of the diode. Thus again power consumption and loss are increased.
This invention relates to improvements to some of the devices described above and to solutions to the problems raised or not solved thereby.